Polyimide-Based Flexible Coupled-Coils Design and Load-Shift Keying Analysis
PPolyimide-Based Flexible Coupled-Coils Design and Load-Shift Keying Analysis
Yuan Yao, Wing-Hung Ki and Chi-Ying Tsui
Department of Electronic and Computer Engineering The Hong Kong University of Science and Technology Hong Kong SAR, P. R. China [email protected], [email protected], [email protected]
Abstract β Wireless power transfer using inductive coupling is commonly used for medical implantable device. The design of the secondary coil on the implantable device is important as it will affect the power transfer efficiency, the size of the implant and also the data transmission between the implant and the in-vitro controller. In this paper, we present a design of the secondary coil on polyimide-based flexible substrate to achieve high power transfer efficiency. Load shift keying modulation is used for the data communication between the primary and secondary coils. Thorough analysis is done for the ideal and practical scenario and it shows that a mismatched secondary LC tank will affect the communication range and communication correctnes. A solution to achieve robust data transmission is proposed and then verified by SPICE simulations. KeywordsβImplantable Medical Devices; Flexible inductive coil; Load-Shift Keying. I. I NTRODUCTION
Implantable medical devices (IMD) usually have a high constraint on the size, and thus wireless power transfer is widely used instead of the bulky batteries to provide power for the implants. A wirelessly powered trans-sclera electrical stimulation (TsES) system has been developed to treat retinal degenerative diseases [1]. However, the power transfer efficiency of the inductive link was only 8% at a distance of 10 mm, which may not adequate for in vivo operations. The secondary coil was fabricated on a printed circuit board (PCB), which was too rigid to be used inside the body. Moreover, implant biocompatibility could not be guaranteed. In this paper, to solve the above problem, a pair of polyimide-based flexible coupled-coils is proposed to solve the above problems. To save the use of an additional antenna, power and data are transmitted via the same inductive coupling link. Frequency-shift keying (FSK) modulation is commonly used in IMDs applications to achieve high data rate [2], but using FSK will increase circuit complexity, and high data rate is not necessary for many applications. To minimize the system volume, load-shift keying (LSK) modulation is also commonly used [3]-[5]. However, it was found that the uplink data sent back from the implant to the transmitter is not as robust as expected [1], limiting the communication range. In this paper, the issue of using LSK scheme is studied and analyzed. A solution is then suggested and verified by SPICE simulations. II. P OLYIMIDE -B ASED F LEXIBLE C OUPLED -C OILS D ESIGN
IMDs are usually low-power applications and inductive power transfer using a pair of series-parallel (S-P) resonant coupled-coils is usually used. In [1] and [6], for an trans-sclera electrical stimulation system, the secondary coil is fabricated on a PCB with thickness of 0.3 mm, and after packaging with encapsulation, the implant thickness will be up to 1 mm. Surgeons suggest to put it beneath the conjunctiva and rectus muscle, and the implant width should be less than 6 mm so as not to affect the activity of the eyeball. Polyimide has been proven to be a biocompatible material and an excellent choice for neuroprosthetic application [7], [8]. In this research the polyimide-based flexible secondary coil has been developed to overcome the size limitation and (a) (b) Fig. 1. Proposed secondary coil installation [9]
Table I Proposed Coupled-Coils Parameters
Primary Coil Secondary Coil
Outer Length π N/A 14 mm Self-inductance πΏ nH nH Series resistance π π @ 40.68 ππ»π§ (a) (b) Fig. 2. Polyimide-based flexible coupled coils simulation setup (a) flat case. (b) bending case mprove the power transfer efficiency (PTE). The proposed secondary coil installation is shown in Fig. 1. The two electrodes and coil are fabricated on the polyimide substrate, and the controller IC was mounted on the substrate as shown in Fig. 1. Proposed secondary coil installation [9] . As the substrate is biocompatible, the biocompatible packaging is easier to achieve, and the implant thickness is reduced to 0.2 mm. With the flexible substrate, the implant can be attached to the surface of the eyeball so that the implant can be made larger without affecting the eyeball movement. The width is still kept at 6 mm, while the length is increased to 40 mm for better stimulation performance. The shape of the coil is oval to make better use of the area. To increase the link efficiency, the primary coil could also be made on a flexible PCB, such that it can be attached onto the skin to minimize the distance between the coils. The secondary coil is bended and attached to the surface of the eyeball. Bending will change both the self-inductances and the coupling coefficient, and the field solver HFSS is used to simulate the coupled-coils under the bending situation. The simulation setup is shown in Fig. 2. Parameters of the coupled-coils are shown in Table I. The primary has series resonance and the secondary has parallel resonance. The inductive link circuit model is shown in Fig. 3. From [10], the inductor and capacitor values under the resonant condition is related by the following equation: and the PTE is given by When both the primary and secondary were flat and 10 mm apart, the simulated PTE of the inductive link reached 39.33%. When both coils were bended, the self-inductance decreased to 562 nH and the coupling coefficient changed to 4.2% from 5.0% and the PTE dropped to 37%. The flexible coupled-coils can maintain high PTE even when bended. πΏ π 2 is 562 nH, and the resonant capacitor C s thus equals to 27 pF, which is fabricated on-chip. The on-chip capacitance C s cannot be fabricated with a very exact value and at the same time, is affected by the parasitic capacitor of the loading circuits. One effect of the mismatched C s is that the LSK scheme may malfunction at the weak coupling condition and robust data transmission will be compromised. In Section III, the issue of LSK scheme is analyzed and a solution is proposed to achieve robust communication. III. A NALYSIS OF
LSK M ODULATION S CHEME A. Ideal LSK Modulation Scheme
We first consider the LSK modulation under ideal situation. Fig. 4 shows an AC voltage source driving the primary LC tank πΏ π 1 and πΆ π 1 . The secondary LC tank receives the coupled power and delivers to the implant unit. The equivalent loading resistance π πΏπππ is 12.5 kΞ©, and parasitic capacitor πΆ π2 is assumed to be small and can be ignored. During backscattering, the implant unit shunts π ππ to modify the loading resistance across the secondary coil. The primary coil input current is given by (4). The primary impedance π πππ and the primary resonator impedance π were defined on (5) respectively. The equivalent impedance model is shown in Fig. 4(b), and the equivalent impedance π ππ (ππ) is given by π ππ is real at the frequency π π . For low-power application, the load resistor π πΏ would be large, for example, π πΏ > 100 πΊ or even larger than 1 kΞ©, so that the factor πΏ π 2 /πΆ π 2 π πΏ2 is much smaller than 1. As discussed in [11], there is only minor advantage in making the equivalent impedance real, while the analysis is made much simpler with good accuracy by designing πΏ π 2 and πΆ π 2 to satisfy π π = π 2 πΆ π 2 . We further assume that the output voltage is adequately filtered, which requires π π πΆ π 2 π πΏ β« 1 . Moreover, R L is much larger than the parasitic resistor π π 2 . Hence, the imaginary C s1 L s1 I L s2 R s2 C s2 R SW R L kV s R s1 C p2 I (a) C s1 Ls1I ( )L s2 C s2 R SW R L V s I Z eq knV +- R s1 R s2 +- V +- V C p2 (b) Fig. 4. (a) Simplified block diagram of IMDs. (b) Reflected impedance model πΌ (ππ) = π π π πππ (4) π πππ = π + π ππ , π = π π 1 + πππΏ π 1 + π 1 (5) π ππ (ππ) = π π π π 2 + πππΏ π 2 + 1/πππΆ π 2 ||π πΏ (6) π π = 1βπΏ π 2 πΆ π 2 β1 β πΏ π 2 πΆ π 2 π πΏ2 (1) C s1 L S1 R s1 I L S2 R s2 C s2 I R L kV s Fig. 3. Circuit model of inductive link π = π ππ’π‘ π ππ = π π π (1 + π πΌ + π π π )(πΌ + π ) (2) π = π π πΏ π 1 π π 1 , π = π π πΏ π 2 π π 2 , πΌ β‘ π π πΆ π 2 π πΏ (3) art is negligible compared to the real part R eq , and R eq is given by According to (9) R eq increases monotonously w.r.t. R L . Ideally, backscattering changes R eq from π πΏ to π πΏ ||π ππ . The equivalent impedance π ππ and π πππ will decrease so πΌ will increase. By detecting the amplitude of πΌ , the transmitter side can decode the uplink signal. B. The Secondary LC Tank Mismatch Effect
In practice, it is difficult to make πΏ π 2 , πΆ π 2 and R L to satisfy (1). The receiver circuits such as the over-voltage protection (OVP) block and the rectifier will introduce parasitic capacitance C p to the secondary LC tank. The OVP circuit needs large diodes to sink excessive power, and the rectifier diodes have to be large enough to reduce conduction loss, and both together lead to a large C p . The secondary coil is designed to have high quality factor Q to improve PTE, and πΏ π 2 is designed to be 564 nH and πΆ π 2 to be 27 pF. From actual measurement, the parasitic capacitance C p can be up to 12 pF [1], which is not negligible and affects the equivalent impedance. The equivalent impedance Z eq ( s ) is shown in (10). πΏ π 2 , πΆ π 2 and R L cannot satisfy (1) due to C p . Hence, π ππ is not monotonously changing with π πΏ . Moreover, the primary impedance π πππ is not monotonously increasing with π πΏ . During backscattering, the secondary coil switches from heavy load to light load and the change in the primary impedance Ξπ πππ is given by (7): Without the parasitic capacitor, Ξπ πππ is a monotonous function of π πΏ , and Ξπ πππ is positive under all coupling condition. However, due to C p of the secondary LC tank, π πππ π» will be smaller than π πππ πΏ at the weak coupling condition. In the numerical example, the parameter set up were following the structure shown in Fig. 4(a), πΏ π 1 , πΏ π 2 , π π 1 and π π 2 were set according to Table I. πΆ π 1 and πΆ π 2 were set to satisfy (1). π πΏ was 12.5 kΞ© and π ππ was set to 500 Ξ©. As shown in Fig. 5 , when k is lower than 0.05, Ξπ πππ is negative, which means that during backscattering when the secondary loading is switched from light load to heavy load, the amplitude of the primary coil current πΌ will decrease instead of increasing, and according to (8) ΞπΌ is positive at weak coupling. The theoretical results were shown in Fig. 6. With the coupling coefficient increasing, ΞπΌ will change from positive to negative, so the uplink signal generated by the LSK scheme will flip consequently. The situation will be worse if there is also a mismatch in the primary LC tank. We change the simulation set up. The primary resonant capacitance πΆ π 1 is larger than the designed value by 1% and πΆ π is still 12 pF. The results of the stimulation are shown in Fig. 5 and Fig. 6 , which are labeled as βPrimary mismatchβ. From the stimulation results, it can be seen that due to the primary mismatch, the uplink signal will flip at k = 0.09 instead of 0.05, and hence the effective communication distance will further decrease. π ππ = π π πΏ π 1 πΏ π 2 β π πΏ + π π 2 + π β πΆ π πΏ2 π π 2 β ππ(πΏ π 2 β πΆ π πΏ2 + π πΆ πΏ π 2 π πΏ2 )(π πΏ + π π 2 ) + π [πΏ π 22 + π πΏ2 πΆ (π π 22 πΆ β 2πΏ π 2 )] , πΆ = πΆ π 2 + πΆ π (10) π ππ β π π2 π πΏ π 1 πΏ π 2 (1 + π π πΏ )π π 2 , π πΏ = π π πΏ π 2 π πΏ (9) Fig. 5. Primary impedance magnitude difference versus coupling coefficient Ξπ πππ = π πππ πππβπ‘ β π πππ βπππ£π¦ (7) ΞπΌ = πΌ πππβπ‘ β πΌ βπππ£π¦ (8) Fig. 6. Primary resonator input current magnitude difference versus coupling coefficient
Fig. 7. Detuned primary LC tank input current magnitude difference versus coupling coefficient V. S IMULATION R ESULTS
The theoretical results are shown in Fig. 7. Two cases were simulated by using SPICE to verify the theoretical results. For both cases, πΆ π was set to 12 pF, and k at 0.06. For the proposed solution, πΆ π 1 is set at 17.03 pF, which is smaller than the designed value by 1%. The switch was driven by the βSWβ signal, and it is active high. When the switch turns on, the secondary loading will change from light load to heavy load, the primary current πΌ should increase but as shown in Fig. 8(a), πΌ was larger when the secondary was at light load. The flipping issue has been eliminated in Fig. 8(b). The secondary parasitic capacitance πΆ π will introduce an negative imaginary part into the equivalent impedance π ππ . However, if πΆ π 1 is made larger than the designed value, π will contain a positive imaginary part that could be used to cancel the imaginary part of π ππ , so that Ξπ πππ may become negative at weak coupling. To avoid cancelling between π ππ and π , the primary resonator could be detuned. By making πΆ π 1 lower than the designed value, π was guaranteed to have a negative imaginary part, such that the uplink signal flipping issue could be eliminated. V. C ONCLUSIONS
In this work, a polyimide-based flexible coupled-coils was developed for the TsES system, and the inductive link has been modeled in HFSS and fabricated by FPC. The PTE was 39% when the coils were separated by 10 mm. LSK scheme with parasitic parameter has been analyzed and a solution was proposed. A
CKNOWLEDGMENT
This research is in part sponsored by β¦. R
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Table II Inductive Links Comparison [1] [12] [13] This work
Link structure 2-coil 2-coil 2-coil 2-coil Substrate material FR4 FR4 PDMS Polyimide Operation frequency( Hz ) 40.68 M 5 M 4 M 40.68 M Relative distance( mm ) 10 10 12 10 Outer diameter of primary coil( mm ) 20 70 37 24 Outer diameter of Secondary coil( mm ) 4.4 20 16Γ10 6Γ14 Distance/secondary coil diameter 2.3 0.5 1 1.6 Sim. or Meas. Meas. Meas. Meas. Sim. PTE(@ perfect alignment) 8% 85.8% 21% 39%) 4.4 20 16Γ10 6Γ14 Distance/secondary coil diameter 2.3 0.5 1 1.6 Sim. or Meas. Meas. Meas. Meas. Sim. PTE(@ perfect alignment) 8% 85.8% 21% 39%